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<title>The Class-A Amplifier Site - Modular Pre-Amplifier Design</title>
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<body background="slate.jpg" bgcolor="#E8E8E8" lang=EN-GB link=blue vlink=blue>
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<div class=Section1>
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<p align=right style='text-align:right'><b>The Class-A Amplifier Site</b></p>
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<p align=right style='text-align:right'><span style='font-size:10.0pt'>This
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page was last updated on 13 January 2002</span></p>
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<p><b><span style='color:blue'>[</span> <a href="index-2.htm"
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title=index-2.htm>Back to Index</a> <span style='color:blue'>]</span></b></p>
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<p> </p>
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<p align=center style='text-align:center'><b><span style='font-size:20.0pt'>Modular
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Pre-Amplifier Design</span></b></p>
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<p align=center style='text-align:center'>(Wireless World, July 1969)</p>
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<p> </p>
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<p align=center style='text-align:center'><b>Optimally designed stages that may be used separately or in several different combinations</b></p>
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<p> </p>
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<p align=center style='text-align:center'><b>by J. L. Linsley Hood, </b><b><span
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style='font-size:8.0pt'>M.I.E.E.</span></b></p>
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<p align=center style='text-align:center'> </p>
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<p align=center style='text-align:center'> </p>
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<p>The type of distortion introduced by a class A transistor amplifier
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operating at a low signal level will be predominantly second harmonic and
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inoffensive to the ear. Although harmonic distortion is a convenient thing to
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measure, and makes a reasonable yardstick for comparative purposes, at low
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levels its presence is less important than that of the intermodulation effects
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it causes. When a complex signal is transmitted through a non-linear element,
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intermodulation products between the separate components of the signal are
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formed, and these are readily apparent in the final audible result as a
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<EFBFBD>blurring<EFBFBD>, and the loss of separate identity, of the individual components which
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make up the whole. A measure of this is the ease (or difficulty) in
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distinguishing the words of a choral performance in the presence of an
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orchestral background, or in identifying the presence and nature of individual
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instruments in a large orchestra.</p>
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<p> </p>
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<p>Measurements by a number of workers <span style='font-size:8.0pt'>(1)</span>
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have indicated that the magnitude of intermodulation products can be much
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greater than that of the total harmonic distortion level, and the
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non-linearities which are likely to be of the most importance in this respect
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are those at the low- and high-frequency ends of the audible range.</p>
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<p> </p>
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<p>At the moment, the performance of audio amplifiers is much superior in this
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respect to that of f.m. transmissions, tape recordings, disc replay systems, or
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loudspeakers. However, advances in manufacturing techniques of gramophone
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records, pickup cartridges and loudspeakers have allowed a continuing
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improvement in the performance of these in harmonic and i.m. distortion, and it
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is clear that any amplifier design offered at this time should have a very high
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standard of performance if it is to remain of continuing value over the next
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decade.</p>
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<p> </p>
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<p align=center style='margin:0cm;margin-bottom:.0001pt;text-align:center'><a href="jlhmodprefig1a.gif" target="_blank"><span style='color:windowtext;text-decoration:none'>
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<img border=1 width=708 height=268 src="jlhmodprefig1.gif"></span></a></p>
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<p align=center style='text-align:center'>Fig. 1. A likely combination of stages.</p>
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<p align=center style='text-align:center'><span style='font-size:10.0pt'>(Click on figure for a higher resolution image)</span></p>
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<p> </p>
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<p>The author has designed a range of high-quality pre-amplifier stages. Each
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stage performs its required operation with negligible noise and distortion.
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When joined together, as for example in Fig. 1, the total harmonic distortion
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level is below 0.1% over the frequency range 20Hz-20kHz, at any tone control
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setting, and for up to 2V r.m.s. output. Each stage is capable of operating on
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its own and has an output impedance low enough for screened cable
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inter-connections to be made without high frequency loss.</p>
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<p> </p>
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<p><b>Magnetic pickup equalization circuit</b></p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=500 height=310 src="jlhmodprefig2.gif"></p>
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<p align=center style='text-align:center'>Fig. 2. Phase-inverting amplifier stage used to obtain R.I.A.A. replay characteristic.</p>
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<p> </p>
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<p>The required R.I.A.A. replay characteristics can be approximated by several
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different circuit arrangements. The most straight-forward from the point of
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view of performance calculation is that shown in Fig. 2, employing a simple
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phase-inverting amplifier stage. If the gain of amplifier M is high enough,
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point Z becomes a virtual earth (see Appendix I), and the input impedance of
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circuit equivalent to that of the input network B. The load resistance required
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by the pickup cartridge, usually 47-50kohm, is provided by a suitable choice of
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R1. With resistor R2 equal to R1, stage gain is given by R4 + R5/R5 at the
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mid-point frequency (usually 1kHz) if the impedance of C2 is large, and that of
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C3 small in relation to R2. Since the voltage output to be expected from most
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good quality magnetic pickup cartridges is in the range 4-10mV for a 5cm/sec
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recorded velocity, a gain of 10 is adequate for this stage. The required replay
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frequency-response curve shown in Fig. 3 can be obtained by a suitable choice
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of C2 and C3. Since the two networks A and B determine the frequency response
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of this circuit, it is apparent that substitution of these can be made to
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provide a wide range of different performance characteristics without
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alteration to the circuit of amplifier unit M.</p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=465 height=291 src="jlhmodprefig3.gif"></p>
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<p align=center style='text-align:center'>Fig. 3. Required R.I.A.A. frequency-response curve and circuits approximation to this.</p>
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<p> </p>
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<p>The final circuit can be seen at the front of Fig. 1. Because phase
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inversion between input and output is required, and because the necessary gain
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is higher that can be obtained from any single transistor arrangement, a
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triplet circuit has been used. Tr1 and Tr3 are high-gain, low-noise
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voltage-amplifying stages, and Tr2 is a phase and voltage transformation stage
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allowing the input transistor to be used in its most linear region. The output
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transistor has a low collector load resistance, to reduce distortion to the
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lowest possible level.</p>
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<p> </p>
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<p>D.C. working-point stability is ensured by D.C. negative feedback through R3
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and R2 to the base of Tr1, and through R4 to the emitter circuit of the same
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transistor. The circuit R4, C4, and C5 also provides the feedback path
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necessary, in conjunction with the input capacitor C1, to provide an
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18dB/octave steep-cut rumble filter, with a turn-over frequency of 25Hz (see
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Appendix II), and an ultimate attenuation of more than 40dB at 8Hz.</p>
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<p> </p>
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<p>Capacitor C6 provides phase correction, and is essential for a clean
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square-wave response, and freedom from transient ringing, when used with a
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capacitive load.</p>
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<p> </p>
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<p>The response of this circuit is particularly good, and it can deliver up to
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1 volt output with distortion less than 0.02% from 100Hz to 10kHz.</p>
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<p> </p>
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<p><b>Stages for ceramic cartridge equalization</b></p>
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<p><b> </b></p>
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<p align=center style='text-align:center'><img border=1 width=600 height=396 src="jlhmodprefig4.gif"></p>
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<p align=center style='text-align:center'>Fig. 4. Impedance conversion stage for use with ceramic cartridge. This may be<br />
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directly substituted for the magnetic cartridge stage at the front of Fig. 1.</p>
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<p align=left style='text-align:left'> </p>
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<p>Fig. 4 is an impedance conversion stage contributing less than 0.05%
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distortion at 1kHz and having a flat response from 35Hz to greater than 200kHz,
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with 18dB/octave roll-off below 35Hz. This simple stage may be directly
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substituted for the magnetic cartridge stage of Fig.1.</p>
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<p> </p>
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<p>Alternatively, should it be required that the pre-amplifier be able to cope
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with inputs from both magnetic ceramic cartridges, then switchable equalization
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networks for A and B can be provided. These are shown in Fig. 5. When used with
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a ceramic cartridge the output is from 50 to 200mV. To preserve the required
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shape of the rumble filter characteristic it is necessary to alter the values
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of C4 and C5 from 25uF to 12.5uF. The pre-amp response is then as shown in Fig.
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5, curve 1.</p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=600 height=504 src="jlhmodprefig5.gif"></p>
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<p align=center style='text-align:center'>Fig. 5. Changes in equalization networks A and B of the magnetic cartridge<br />
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input stage allowing direct use of ceramic cartridge. Components for network<br />"A" are the same for the three curves shown.</p>
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<p align=center style='text-align:center'> </p>
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<p>The performance of many ceramic pickup/amplifier combinations is
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disappointing in comparison with that obtainable from a good magnetic cartridge
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with a similar amplifier. This is sometimes due to the mismatching between
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cartridge and amplifier, or through inadequate input impedance provision (in
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the modification shown in Fig. 5 this is 4.4Mohm), or due to the failure of the
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piezoelectric element within the cartridge to provide the required equalization
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for the 12dB fall in voltage output anticipated when a recording having
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R.I.A.A. velocity characteristics is replayed on a displacement sensitive
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device. In the latter case, a very considerable improvement in the relative
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performance of the ceramic cartridge may be obtained by shunting part of the
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input resistor in the input network B by a small capacitor. Curves 2 and 3 in
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Fig. 5 show partial and complete correction respectively.</p>
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<p> </p>
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<p><b>Tone-control stage</b></p>
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<p><b> </b></p>
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<p>The tone-control stage is of conventional type, and uses a negative feedback
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system derived from the design due to Baxandall <span style='font-size:8.0pt'>(2)</span>.
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However, it differs from normal practice in that a junction field-effect
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transistor is used as the active element. Field-effect transistors have both
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lower noise levels and better linearity than bipolar transistors, and in this
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type of circuit the high input impedance results in negligible loading of the
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tone-control network. The stage gain needed in this circuit requires a high
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value drain load resistor, and the f.e.t. must therefore be followed by an
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emitter-follower to provide the low output impedance desired for easy
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interconnection of the separate units.</p>
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<p> </p>
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<p>If the feedback tone-control network is to perform satisfactorily, both the
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input and output impedances seen by the network at its ends must be low in
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relation to the network input impedance when the sliders of the potentiometers
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are at the position nearest to the point being measured. Some form of impedance
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conversion circuit is therefore also needed between the volume control and the
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tone-control circuit. An emitter follower is also used at this point. The
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0.001uF capacitor in the emitter circuit of Tr4 is to avoid the possibility of
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high frequency parasitic oscillation occurring if long screened leads are used
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to connect the base of Tr4 to the volume control.</p>
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<p> </p>
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<p>The input to this section is taken through a switch from the gramophone
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pre-amplifier section, and other inputs provided with preset gain-equalization
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potentiometers. The switch is arranged to earth the inputs not in use, to
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minimize breakthrough between programme channels. </p>
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<p> </p>
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<p>The gain/frequency characteristics of the stage are shown in Fig. 6.</p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=600 height=337 src="jlhmodprefig6.gif"></p>
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<p align=center style='text-align:center'>Fig. 6. Gain/frequency characteristics of tone control stage.</p>
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<p> </p>
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<p><b>Low-pass filter circuit</b></p>
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<p> </p>
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<p>The voltage amplifying stage preceding the main amplifier should include a
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steep-cut low-pass filter that can be set to remove unwanted high frequencies.
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This can be done either by a suitable LCR filter arrangement, or by an active
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filter giving an equivalent performance without the use of inductors. The
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circuit arrangements available for low-pass active filters are shown in Fig. 7.
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(b) is the well known circuit arrangement first employed in an audio amplifier
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design by P. J. Baxandall <span style='font-size:8.0pt'>(3)</span>, and (d) is
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the unity gain rearrangement of this circuit introduced by Sallen and Key <span
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style='font-size:8.0pt'>(4)</span>. The frequency response of all these circuit
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arrangements is similar, <i>mutatis mutandis</i>, to that shown in Fig. 8, and
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the circuit should be preceded or followed by a simple RC filter if the type of
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response shown in the dotted line is required.</p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=400 height=369 src="jlhmodprefig7.gif"></p>
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<p align=center style='text-align:center'>Fig. 7. Circuit arrangements for active low-pass filter design.</p>
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<p align=center style='text-align:center'> </p>
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<p align=center style='text-align:center'><img border=1 width=574 height=336 src="jlhmodprefig8.gif"></p>
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<p align=center style='text-align:center'>Fig. 8. Frequency response of the active filter circuits is 12dB/octave. Preceding</p>
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<p align=center style='text-align:center'>the filter with RC network gives response shown in broken line.</p>
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<p> </p>
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<p>For a given overall stage gain, type (b) gives much better distortion factor
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near the region of cut-off than (a), and (c) is marginally better than (b) when
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used with non-linear amplifier elements. The particular advantage of (c)
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however, is that it can be used conveniently with a very low-distortion
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two-transistor circuit.</p>
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<p> </p>
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<p>The final stage, with the filter circuitry, is shown in Fig. 1. As a matter
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of practical convenience, the component values of this circuit have been chosen
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so that the required low-pass response is obtained when all of the capacitors
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<EFBFBD>Cx<EFBFBD> are of equal value to each other. The frequency response obtained with a
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given value of <20>Cx<43> can be found in Fig. 9. The user can interpolate between
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these to obtain turn-over frequencies at any points to suit his own
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requirements. If a ganged selector switch is employed to give a range of
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turn-over frequencies, the switch arms (moving contacts) should be connected to
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the junction of the resistors in the RC filter and to the 470ohm resistor in
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the main filter network. In Fig. 1 the 0.0047uF capacitor for <20>Cx<43> results in response
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being 3dB down at 18kHz. With good quality programme sources this is a
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recommended capacitor value.</p>
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<p> </p>
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<p align=center style='text-align:center'><img border=1 width=600 height=323 src="jlhmodprefig9.gif"></p>
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<p align=center style='text-align:center'>Fig. 9. Graph and table of turn-over frequencies for different value of <20>Cx<43>.</p>
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<p> </p>
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<p>With capacitors of zero value, the response of the circuit is flat to about
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100kHz. The user should however arrange for the response to fall off above
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25kHz. (It is unlikely that the listener will find anything to gain from the
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parts of the sonic spectrum beyond this point.)</p>
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<p> </p>
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<p>The optimum performance of this particular type of circuit arrangement is
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obtained when the overall gain is about 50 with feedback. A 20-40mV input is
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therefore adequate for this stage for the output voltages required.</p>
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<p> </p>
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<p>The distortion level of this circuit is less than 0.03% at 2 volts r.m.s.
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output or less, at any frequency within the pass band. The output impedance is
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less than 150 ohms over the range from 20Hz to the cut-off frequency selected.</p>
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<p> </p>
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<p>It is convenient, for several reasons, to operate at the 60-100mV level
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through the tone-control stages. At this output voltage level the distortion
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introduced by a RC coupled f.e.t. stage is less than 0.1% even without
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feedback, so that the maximum <20>lift<66> settings of either <20>bass<73> or <20>treble<6C>
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controls cannot give rise to unacceptable levels of distortion. It is also
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large enough for the noise and inevitable 50Hz pickup to be unobtrusive. Some
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attenuation is therefore desirable between the tone control unit and the
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steep-cut filter circuit. This is obtained by the preset 2kohm potentiometer in
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the tone control circuit, which provides a convenient means for setting the
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overall gain of the amplifier system, and also as a coarse <20>balance control<6F> in
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a stereo system. Fine balance between channels is obtained by adjusting the
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100ohm balance potentiometer in the output stage. This alters the stage gain
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over the ratio 6:10. </p>
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<p> </p>
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<p><b>Constructional notes</b></p>
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<p> </p>
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<p>The constructional technique used by the author in building the prototype of
|
|
|
this amplifier is similar to that used in the 10-watt class-A design described
|
|
|
in Wireless World in April 1969, with the separate units laid out in mirror
|
|
|
image form, as a stereo pair on a single 4in X 4<>in s.r.b.p. pin board, Two
|
|
|
units of each type can be accommodated on each board, laid out more or less in
|
|
|
the form of the circuit diagram(or its mirror image).</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>In general, reasonable care should be taken to separate input from output
|
|
|
leads, and where the boards are to be mounted as a group within the same box,
|
|
|
it would be wise to interpose a sheet metal screen between them.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The units are separately coupled by 250uF capacitors from a common 24-volt
|
|
|
line, derived from a zener diode stabilized RC filter power supply. This supply
|
|
|
is separate from the main amplifier, and a 30mA output is ample. Details of a
|
|
|
suitable power supply are given in Fig. 10. The expected working voltage on
|
|
|
each of the unit sub-rails is about 15volts.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=500 height=263 src="jlhmodprefig10.gif"></p>
|
|
|
|
|
|
<p align=center style='text-align:center'>Fig. 10. Suitable power supply for any combination of stages.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>Apart from the input transistor in the gramophone pre-amp unit (Tr1) for
|
|
|
which the BC109 is to be preferred, there is no particular reason why any
|
|
|
modern silicon planar types should not give an indistinguishable performance.
|
|
|
For example, the n-p-n types could be 2N3904, BC107/8/9, 2N3707, or BC184Ls.
|
|
|
Similarly, the p-n-p types could be 2N4058, 2N3906, or BC214Ls.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>Although, in many cases, the use of 1/4 watt resistors is sufficient, it
|
|
|
would probably be found simpler to use 1/2 watt units throughout. 5% tolerance
|
|
|
carbon film resistors are to be preferred.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The author has mounted the gramophone pickup equalization circuit in a
|
|
|
separate small diecast box, immediately under the gramophone turntable unit, so
|
|
|
that the leads from the gramophone are taken at a low impedance from the output
|
|
|
of this unit. This has been very effective in reducing the hum picked up on the
|
|
|
output leads to an imperceptible level.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Appendix l</b></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The use of <20>virtual earth<74> (null seeking) amplifier circuit arrangements is
|
|
|
superficially ill-advised with input elements such as pickup cartridges,
|
|
|
because it appears that as the operating frequency is increased, the input half
|
|
|
of the balancing limbs will also change, with a resultant change in the gain of
|
|
|
the circuit. In particular a magnetic pickup cartridge may have an inductance
|
|
|
of some 300-800mH and the impedance of this will exceed that of the input
|
|
|
circuit in the range 12-20kHz. This should clearly reduce the gain of the
|
|
|
system by reducing the ratio of A to B.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>However, on reflection, it can be seen that the amplifier operates as a null
|
|
|
generating device, sensitive only to the current flowing in the input circuit
|
|
|
to the <20>virtual earth<74>. As the operating frequency increases, so the current
|
|
|
flow through R1 will decrease, but so it would in any case, regardless of the
|
|
|
amplifier, were the element simply connected across network B as the load
|
|
|
recommended by the cartridge manufacturers (at these frequencies the impedance
|
|
|
of C1 can be ignored), and the voltage across R1 measured by a perfect voltage
|
|
|
amplifier. The decrease in current input into a given resistive loads from a
|
|
|
source having a series inductance is simply an unfortunate fact of life, from
|
|
|
which one cannot escape, whatever one<6E>s technique of measurement, and high
|
|
|
impedance voltage amplifiers connected across the load, or low impedance
|
|
|
current amplifiers connected in series with it, are alike in this respect,
|
|
|
except that with transistors, the latter are a bit easier to contrive. The same
|
|
|
argument is also applicable, in the appropriate context, to high impedance
|
|
|
capacitative elements such as piezo-electric pickup cartridges. Once again, the
|
|
|
voltage amplifier and current amplifier see the same phenomena in identical
|
|
|
form. The necessary, and inevitable, corrections can be accomplished by simply
|
|
|
by the tone control settings.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Appendix II</b></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>Although the R.I.A.A. replay characteristics suggest an approximately flat
|
|
|
velocity response from 20-50Hz, this would effectively imply recording bass
|
|
|
lift in this region and the author suspects that this is not done and a
|
|
|
constant modulation characteristic being used instead. The author has
|
|
|
therefore, for his own use, modified the values of the feedback elements as
|
|
|
follows: R5 <20> 470 ohms; R6 <20> 1.5kohms; C1 <20> 0.47uF; C3 <20> 6800pF; and C6 <20>
|
|
|
6800pF. These changes maintain the velocity response flat down to 25Hz, with
|
|
|
rapid attenuation below this frequency. Unfortunately the mid point gain of the
|
|
|
circuit is reduced to 5, and some additional amplification is therefore needed
|
|
|
if it is desired to avoid working with the tone control circuit at the 20mV
|
|
|
level. The simple floating emitter collector-follower circuit of Fig. 11 is
|
|
|
therefore interposed, without coupling capacitors, between the output series
|
|
|
resistor and the collector of Tr3. The distortion contributed by this is less
|
|
|
than 0.05%.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=500 height=378 src="jlhmodprefig11.gif"></p>
|
|
|
|
|
|
<p align=center style='text-align:center'>Fig. 11. Floating emitter collector-follower circuit referred to in Appendix II.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>References</b></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p style='margin-left:32.2pt;text-indent:-18.0pt'>1.<span style='font:7.0pt "Times New Roman"'>
|
|
|
</span>Langford-Smith, F., <20>Radio Designers Handbook<6F>, Vol.4 ch.72.</p>
|
|
|
|
|
|
<p style='margin-left:32.2pt;text-indent:-18.0pt'>2.<span style='font:7.0pt "Times New Roman"'>
|
|
|
</span>Baxandall, P. J., <20>Negative-Feedback Tone Control<6F>, Wireless World,
|
|
|
October 1952</p>
|
|
|
|
|
|
<p style='margin-left:32.2pt;text-indent:-18.0pt'>3.<span style='font:7.0pt "Times New Roman"'>
|
|
|
</span>Baxandall, P. J., <20>Gramophone and Microphone Pre-amplifier<65>, Wireless
|
|
|
World, January 1955</p>
|
|
|
|
|
|
<p style='margin-left:32.2pt;text-indent:-18.0pt'>4.<span style='font:7.0pt "Times New Roman"'>
|
|
|
</span>Sallen, R.P. and Key, E.L., I.R.E. Trans. Circuit Theory, March 1955, p.
|
|
|
74-85</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b><span style='font-size:16.0pt'>Postscript</span> </b>(December 1970)</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Modular pre-amplifier</b></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The intention in the original article was not to offer a complete
|
|
|
pre-amplifier design, but rather to describe a series of versatile <20>building
|
|
|
blocks' from which the potential user could assemble a 'custom built<6C>
|
|
|
pre-amplifier to suit his own needs or preferences. To increase the scope of
|
|
|
this some additional circuit modules are described below.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Steep c</b><b>ut low-pass filter.</b> It is certainly prudent to include
|
|
|
a low-pass filter somewhere fairly close to the input of the main amplifier
|
|
|
whenever a wide-bandwidth main amplifier is to be used with a good-quality loudspeaker
|
|
|
system. Doing so will prevent unwanted high-frequency components, arising from
|
|
|
component noise, record surface noise, and similar causes, from impairing the
|
|
|
long-term listening comfort of the user, and from producing avoid<69>able
|
|
|
intermodulation effects due to non-linearities in the loudspeakers.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The combination of such a steep-cut low-pass filter with a low-distortion,
|
|
|
low-output impedance driver stage, with a gain of 50 and an output capability
|
|
|
of some 2V r.m.s. at 0.02% t.h.d., appeared to provide the most versatile
|
|
|
system for use with a wide variety of power amplifiers.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>However, many power amplifiers require an input voltage of only 0.25 - 0.8V
|
|
|
r.m.s., and there are snags in respect of hum and component noise if the stages
|
|
|
following the volume control are operated at levels below some 50mV. The
|
|
|
preferred level to achieve an optimum balance of noise and distortion
|
|
|
components is probably in the 100 <20> 200mV region. In these circumstances a
|
|
|
driver-stage gain of 50 is excessive, and much of the available gain must be
|
|
|
removed by an input attenuator, and if a potentiometer is used for this it can
|
|
|
introduce noise.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='margin:0cm;margin-bottom:.0001pt;text-align:center'><img border=1 width=746 height=401 src="jlhmodprefigps5.gif"></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>To meet this need more conveniently, two further versions of the driver
|
|
|
amplifier, incorporating steep-cut low-pass filter characteristics which are
|
|
|
identical to that of the original circuit, and having gains of 20 and 5, are
|
|
|
shown in Figs. 5(a) and 5(b). An alternative, three-transistor arrangement
|
|
|
whose cut-off slope is variable over the range <20>6 to <20>18dB/octave, at any
|
|
|
chosen (switchable) frequency, is shown in Fig. 6. This consists of a single
|
|
|
transistor version of the <20>H<EFBFBD> filter used in the two previous pre-amplifier
|
|
|
designs (the nomenclature derives from the shape of the component layout in the
|
|
|
<EFBFBD>op-amp<6D> form), followed by a very low distortion two-transistor amplifier
|
|
|
whose gain can be chosen as required, over the range 5 to 100, by adjustment of
|
|
|
Ra and Rb. If a unity-gain stage is all that is required (actually the gain is
|
|
|
about 0.9) the output ran be taken from the point marked 'A' on the diagram,
|
|
|
and Tr7 and Tr8 omitted.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=456 height=600 src="jlhmodprefigps6.gif"></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The response curve of the filter circuit, at any chosen turnover frequency
|
|
|
is shown in Fig. 7. The slope is smoothly variable by adjustment to the 5kohm
|
|
|
pot. If the slope pot. Is open circuit the response is flat to 20kHz and
|
|
|
beyond, but in this case the load impedance should not be less than 50kohm.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=394 height=249 src="jlhmodprefigps7.gif"></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>For completeness, an equivalent single-transistor high-pass filter, having a
|
|
|
cut-off slope approaching 18dB/octave, and suitable for use as a <20>rumble<6C>
|
|
|
filter or a pre-amplifier woofer/tweeter cross-over filter, is shown in Fig. 8.
|
|
|
The frequency response characteristics of this filter are shown in Fig. 9. Both
|
|
|
of these filter circuits should be driven from a source having a fairly low
|
|
|
impedance <20> not higher than 6kohm.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=388 height=265 src="jlhmodprefigps8.gif"></p>
|
|
|
|
|
|
<p align=center style='text-align:center'> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=420 height=301 src="jlhmodprefigps9.gif"></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>If single transistor <20>H<EFBFBD> filters are to be used at output signal levels
|
|
|
exceeding 100mV a Darlington transistor, e.g. Motorola MPSA14, is to be
|
|
|
preferred.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The apparent noise level, referred to the input, of the two transistor
|
|
|
driver amplifiers, using reasonably low noise transistors and an input
|
|
|
impedance of the order provided in the normal circuit, is about 4 <20> 6uV. The
|
|
|
output noise voltage in the original circuit was 0.2 <20> 0.3mV, which should be
|
|
|
inoffensive. With a lower gain driver stage this noise will be reduced even
|
|
|
further.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The use of a variable negative feedback type of balance control in these
|
|
|
circuits is deliberate, in that it permits a low output impedance to be
|
|
|
obtained from the driver stage. Measurements made with a wide range of
|
|
|
published transistor-operated power amplifiers have shown that substantially
|
|
|
lower distortion levels are often given by using a low-impedance drive circuit,
|
|
|
and that there is frequently an advantage also in terms of hum, noise, and
|
|
|
transient response.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Tone-control circuit.</b> This stage has a worst case (bass and treble
|
|
|
controls set to maximum <20>lift<66>) distortion level which is typically less than
|
|
|
0.1% at 1V r.m.s. output. It is perfectly capable of driving a normal high-quality
|
|
|
power amplifier without the interposition of other pre-amplifier stages. The
|
|
|
required signal amplification could then be provided prior to the volume
|
|
|
control. This is tending to be the normal practice in commercial <20>hi-fi<66>
|
|
|
amplifiers, in that it gives the highly-sought-after zero noise-level at
|
|
|
minimum volume control settings, and makes for economies in the use of
|
|
|
components.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>Noise in the tone-control stage due to the f.e.t. has caused occasional
|
|
|
troubles. This should not occur with the f.e.t. now recommended for this part
|
|
|
of the circuit (the Amelco 2N4302), which appears to have a consistently low
|
|
|
noise level. The necessary bias adjustments were described in a letter to the
|
|
|
editor published in April 1970.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The input impedance level suggested for the tone-control stage was 50kohm,
|
|
|
because it was thought that most of the other systems likely to be used with
|
|
|
this unit would be transistor operated; and this would be a suitable level for
|
|
|
this purpose, while avoiding some of the hum pick-up problems likely to be
|
|
|
encountered at higher impedance levels. However, if this impedance is too low,
|
|
|
and if a high gain (beta greater than 400) transistor is selected for Tr4 <20> in
|
|
|
fact most BC109s will do <20> the base bias resistors can be increased to 1Mohm
|
|
|
and 560kohm (instead of 200kohm and 100kohm) enabling the volume control and
|
|
|
auxiliary control potentiometers to be increased to 25kohm.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>If an even higher input impedance is required, the f.e.t. impedance
|
|
|
conversion shown in Fig. 4 in the original pre-amp article can be substituted
|
|
|
in its entirety for Tr4. To preserve the function of the rumble filter in this
|
|
|
circuit, with the 0.47uF capacitor desired to feed the tone-control network, a
|
|
|
4.7kohm resistor should be connected from the output side of this capacitor to
|
|
|
the earth line. A low noise f.e.t. is of course preferable.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>If additional amplification is required on any signal source prior to the
|
|
|
tone-control stage (if this is working at the 100mV level) a simple
|
|
|
single-transistor feedback amplifier such as that shown in Fig. 10, can be used
|
|
|
with confidence, in that its performance is stable, its noise is low, it is
|
|
|
almost impossible to damage by an input overload, and its distortion is well
|
|
|
below 0.1% at output voltages up to 0.25V r.m.s., and with gains up to 10.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p align=center style='text-align:center'><img border=1 width=392 height=307 src="jlhmodprefigps10.gif"></p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Magnetic pickup equalisation circuit.</b> Some requests have been
|
|
|
received for component values for the use of this circuit for tape-replay
|
|
|
characteristic equalization. The author remains of the opinion that this type
|
|
|
of provision is best left to the manufacturers of the tape recorder, in that
|
|
|
the actual head characteristics can influence the replay frequency/voltage
|
|
|
characteristics.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>However, a fairly close approximation to the replay curve theoretically
|
|
|
required for 7.5 i.p.s. is given if C2 and R2 in the original equalization
|
|
|
network A are altered to I00pF and 27kohm.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The noise level of this circuit is almost entirely determined by the performance
|
|
|
of Tr1 The BC184C and 2N5089 transistor types may be of interest in this
|
|
|
position.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>The maximum output which can be obtained from this circuit at 0.02% t.h.d.,
|
|
|
is 2V r.m.s. If the normal input to the tone control circuit, or other
|
|
|
following stages, is l00mV, this gives a 26dB overload capability. The gain of
|
|
|
the equalization circuit can be increased by a factor of 3, (i.e. to 30 at
|
|
|
1kHz) without upsetting the rumble filter characteristics if R5 is reduced to
|
|
|
68ohm and C4 increased to I00uF.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p><b>Miscellaneous.</b> An omission from the original article was the
|
|
|
suggestion that high value resistors (2 <20> 5Mohm) should be connected across the
|
|
|
switch contacts, from slider to each Cx. This removes 'plops' on switching
|
|
|
ranges.</p>
|
|
|
|
|
|
<p> </p>
|
|
|
|
|
|
<p>A number of correspondents have queried the need for a separate h.t. power
|
|
|
supply for the pre-amp. (The reservoir capacitors for the unit shown should
|
|
|
have read 35V working, not 25V). It is always possible to run the pre-amp via a
|
|
|
suitable voltage-dropper circuit from the main amplifier power supply and if a
|
|
|
zener diode is included in this line, this scheme may be satisfactory. However,
|
|
|
measurements on channel separation and harmonic and i.m. distortion, with
|
|
|
identical amplifier systems invariably show some advantage, particularly at the
|
|
|
low-frequency end of the audible spectrum, in the use of a separate power
|
|
|
supply for the pre -amp (even when the electrolytic bypass capacitors are still
|
|
|
new) and this arrangement is still recommended by the author as well worth the
|
|
|
small additional cost.</p>
|
|
|
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<p> </p>
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<p>One point which has not been published, to the best of the author's
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knowledge, concerns the particular advantage conferred by the feedback pair
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amplifier using complementary transistors, such as that used in the low-pass
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filter circuit, in comparison with the more usual n-p-n/n-p-n pair, where the
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bias for the first transistor is derived from the h.t. line. In the case of the
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n-p-n/p-n-p pair, any h.t. line feedback, due to inadequate h.t. line bypass,
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will be negative rather than positive, and this can assist in obtaining good
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t.h.d. figures down to low signal frequencies.</p>
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<p> </p>
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<p> </p>
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<p> </p>
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<p>My<EFBFBD> thanks go to Malcolm Jenkins for providing the copy of the article used
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to make this web page and to Lynn Miller for converting it into web format.</p>
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<p> </p>
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<p><span style='font-size:8.0pt'> </span></p>
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<p><b><span style='color:blue'>[</span> <a href="index-2.htm"
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title=index-2.htm>Back to Index</a> <span style='color:blue'>]</span></b></p>
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<p> </p>
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<p> </p>
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<p><span style='font-size:8.0pt'><b>HISTORY:</b> Page created 06/01/2002 </span></p>
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<p><span style='font-size:8.0pt'>13/01/2002 Hi-res Fig. 1 and December 1970 postscript added</span></p>
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